Method and Apparatus for Determining a Target Light Intensity From a Phase-Control Signal

ABSTRACT

A dimmable ballast circuit for a compact fluorescent lamp controls the intensity of a lamp tube in response to a phase-control voltage received from a dimmer switch. The ballast circuit comprises a phase-control-to-DC converter circuit that receives the phase-control voltage, which is characterized by a duty cycle defining a target intensity of the lamp tube, and generates a DC voltage representative of the duty cycle of the phase-control voltage. Changes in the duty cycle of the phase-control voltage that are below a threshold amount are filtered out by the converter circuit, while intentional changes in the duty cycle of the phase-control voltage are reflected in changes in the target intensity level and thereby the intensity level of the lamp tube.

CROSS REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. Ser. No. 16/119,670, filed onAug. 31, 2018, which is a continuation of U.S. Ser. No. 15/783,604,filed Oct. 13, 2017, now U.S. Pat. No. 10,070,507, issued Sep. 4, 2018,which is a continuation of U.S. Ser. No. 15/077,090, filed Mar. 22,2016, now U.S. Pat. No. 9,795,019, issued Oct. 17, 2017, which is acontinuation of U.S. Ser. No. 14/323,495, filed Jul. 3, 2014, now U.S.Pat. No. 9,326,356, issued Apr. 26, 2016, which is a divisionalapplication of Ser. No. 13/464,358, filed May 4, 2012, now U.S. Pat. No.8,803,432, issued Aug. 12, 2014, which is a non-provisional applicationof commonly-assigned U.S. Provisional Patent Application No. 61/484,481,filed May 10, 2011, the entire disclosures of which are herebyincorporated by reference.

BACKGROUND OF THE INVENTION Field of the Invention

The present invention relates to a load control device for controllingthe amount of power delivered to an electrical load, such as a lightingload, and more particularly, a phase-control-to-DC converter circuit fordetermining a target intensity level for the lighting load from aphase-control signal.

Description of the Related Art

In order to save energy, high-efficiency lighting loads, such as, forexample, compact fluorescent lamps (CFLs) and light-emitting diode (LED)light sources, are being used in place of or as replacements forconventional incandescent or halogen lamps. High-efficiency lightsources typically consume less power and provide longer operationallives as compared to incandescent and halogen lamps. FIG. 1 is asimplified block diagram of a prior art lighting control system 10having a screw-in compact fluorescent lamp 20. The screw-in compactfluorescent lamp 20 comprises a fluorescent lamp tube 22, which may beformed in a spiral (as shown in FIG. 1). The screw-in compactfluorescent lamp 20 also comprises an enclosure 24 for housing a loadregulation circuit 40 (FIG. 2), e.g., an electronic ballast circuit,which is electrically coupled to the lamp tube 22 for illuminating thelamp tube. The screw-in compact fluorescent lamp 20 has a screw-in base26 adapted to be coupled to a standard Edison socket. The lamp tube 22of a typical prior art screw-in compact fluorescent lamp 20 is filledwith a fill gas of 100% argon at a pressure of approximately 4 Torr.

The lighting control system 10 also comprises a “two-wire” dimmer switch30, which is coupled in series between an alternating-current (AC) powersource 15 and the screw-in compact fluorescent lamp 20 for controllingthe intensity of the lamp tube 22. The dimmer switch 30 may be adaptedto be mounted to a standard electrical wallbox and comprises a hotterminal H coupled to the AC power source 15 for receiving an AC mainsline voltage V_(AC), and a dimmed-hot terminal DH coupled to thescrew-in compact fluorescent lamp 20. The dimmer switch 30 does notrequire a direct connection to the neutral side N of the AC power source15. Examples of prior art dimmer switches are described in greaterdetail is commonly-assigned U.S. Pat. No. 5,248,919, issued Sep. 29,1993, entitled LIGHTING CONTROL DEVICE; U.S. Pat. No. 6,969,959, issuedNov. 29, 2005, entitled ELECTRONIC CONTROL SYSTEMS AND METHODS; and U.S.Pat. No. 7,687,940, issued Mar. 30, 2010, entitled DIMMER SWITCH FOR USEWITH LIGHTING CIRCUITS HAVING THREE-WAY SWITCHES, the entire disclosuresof which are hereby incorporated by reference.

The dimmer switch 30 comprises a bidirectional semiconductor switch 32coupled between the hot terminal H and the dimmed hot terminal DH forconducting a load current LOAD through the screw-in compact fluorescentlamp 20. The bidirectional semiconductor switch 32 may comprise a singledevice, such as a triac, or a combination of devices, such as, twofield-effect transistors (FETs) or insulated-gate bipolar junctiontransistors (IGBTs) coupled in anti-series connection. To control theamount of power delivered to the lamp tube 22, the bidirectionalsemiconductor switch 32 is controlled to be conductive andnon-conductive for portions of a half-cycle of the AC power source 15,such that the bidirectional semiconductor switch is rendered conductivefor a conduction time T_(ON) each half-cycle. The dimmer switch 30 maycomprise a toggle actuator for turning the high-efficiency lamp tube 22on and off and an intensity adjustment actuator for adjusting theintensity of the lamp tube 22 between a low-end intensity and a high-endintensity.

The dimmer switch 30 further comprises a control circuit 34 coupled inparallel with the bidirectional semiconductor switch 32 for conducting acontrol circuit I_(CNTL) through the screw-in compact fluorescent lamp20. The control circuit 34 is coupled to a control input of thebidirectional semiconductor switch 32 for rendering the bidirectionalsemiconductor switch conductive and non-conductive to generate aphase-control voltage V_(PC) using either the forward phase-controldimming technique or the reverse phase-control technique. Accordingly,the bidirectional semiconductor switch 32 is rendered conductive for theconduction time T_(CON) each half-cycle, thus setting a duty cycleDC_(PC) of the phase-control voltage V_(PC). The control circuit 34 maybe operable to provide, for example, a constant gate drive to thebidirectional semiconductor switch 32, such that the bidirectionalsemiconductor switch will remain conductive independent of the magnitudeof the load current I_(LOAD) conducted through the bidirectionalsemiconductor switch and the screw-in compact fluorescent lamp 20. Anexample of a two-wire dimmer switch having a constant gate drive controlcircuit is described in greater detail in commonly-assigned, co-pendingU.S. patent application Ser. No. 12/952,920, filed Nov. 23, 2010,entitled TWO-WIRE DIMMER SWITCH FOR LOW-POWER LOADS, the entiredisclosure of which is hereby incorporated by reference.

The screw-in base 26 of the compact fluorescent lamp 20 provides foronly two electrical connections: a phase-control connection PC to thedimmer switch 30 for receiving the phase-control voltage V_(PC) and aneutral connection NC to the neutral side N of the AC power source 15.The load regulation circuit 40 of the screw-in compact fluorescent lamp20 is operable to adjust the intensity of the lamp tube 22 between thelow-end intensity and the high-end intensity in response to theduty-cycle DC_(PC) of the phase-control signal V_(PC) (i.e., theconduction time of the bidirectional semiconductor switch 32 of thedimmer switch 30).

With forward phase-control dimming, the bidirectional semiconductorswitch 32 is rendered conductive at some point within each AC linevoltage half-cycle and remains conductive until approximately the nextvoltage zero-crossing, such that the bidirectional semiconductor switchis conductive for the conduction time each half-cycle. A zero-crossingis defined as the time at which the AC line voltage transitions frompositive to negative polarity, or from negative to positive polarity, atthe beginning of each half-cycle. Forward phase-control dimming is oftenused to control energy delivered to a resistive or inductive load, whichmay include, for example, an incandescent lamp or a magnetic low-voltagetransformer. The bidirectional semiconductor switch of a forwardphase-control dimmer switch is typically implemented as a thyristor,such as a triac or two silicon-controlled rectifiers (SCRs) coupled inanti-parallel connection, since a thyristor becomes non-conductive whenthe magnitude of the current conducted through the thyristor decreasesto approximately zero amps. Thyristors are typically characterized by arated latching current and a rated holding current, and comprise twomain terminals and a control terminal. The current conducted through themain terminals of the thyristor must exceed the latching current for thethyristor to become fully conductive. In addition, the magnitude of theload current LOAD conducted through the main terminals of the thyristormust remain above the holding current for the thyristor to remain infull conduction.

The control circuits of many forward phase-control dimmers compriseanalog control circuits (such as timing circuits) for controlling whenthe thyristor is rendered conductive each half-cycle of the AC powersource. The analog control circuit typically comprises a potentiometer,which may be adjusted in response to a user input provided from, forexample, a linear slider control or a rotary knob in order to controlthe amount of power delivered to the lighting load. The analog controlcircuit is typically coupled in parallel with the thyristor and conductsa small timing current through the lighting load when the thyristor isnon-conductive.

With reverse phase-control dimming, the bidirectional semiconductorswitch 32 is rendered conductive at the zero-crossing of the AC linevoltage and rendered non-conductive at some point within each half-cycleof the AC line voltage, such that the bidirectional semiconductor switchis conductive for a conduction time each half-cycle. The bidirectionalsemiconductor switch of reverse phase-control dimmers typicallycomprises two field-effect transistors (FETs) in anti-serial connection,or the like. Accordingly, prior art reverse phase-control dimmerswitches have required advanced control circuits (such asmicroprocessors) for controlling the operation of the FETs, and powersupplies for powering the microprocessors. In order to properly charge,the power supply of such a two-wire dimmer switch must develop an amountof voltage across the power supply and must conduct a charging currentfrom the AC power source through the electrical load, in many instanceseven when the lighting load is off.

FIG. 2 is a simplified block diagram of the load regulation circuit 40of the prior art screw-in compact fluorescent lamp 20. The loadregulation circuit 40 comprises an electromagnetic interference (EMI)filter 50 for preventing noise generated by the load regulation circuitfrom being conducted on the AC mains wiring. A full-wave bridgerectifier 52 receives the phase-control voltage V_(PC) from the EMIfilter 50 and generates a rectified voltage V_(RECT). The rectifiedvoltage V_(RECT) is coupled to a bus capacitor C_(BUS) through a diodeD54 for generating a direct-current (DC) bus voltage V_(BUS) across thebus capacitor. The load regulation circuit 40 further comprises aninverter circuit 56 for generating a high-frequency square-wave voltageV_(SQ) from the rectified voltage V_(RECT), and a resonant tank circuit58 for receiving the square-wave voltage V_(SQ) and producing asubstantially sinusoidal high-frequency AC voltage V_(LAMP) (i.e., anarc voltage or lamp voltage), which is provided to the lamp tube 22. Theinverter circuit 56 adjusts the operating frequency f_(OP) of thesquare-wave voltage V_(SQ) in order to adjust the intensity of the lamptube 22.

The load regulation circuit 40 further comprises a phase-to-DC convertercircuit 60 for receiving the rectified voltage V_(RECT) and generating aDC voltage V_(SQ) that has a magnitude that is representative of theduty-cycle DC_(PC) of the phase-control signal V_(PC), and a lampcurrent sense circuit 62 that generates a lamp current control signalV_(ILAMP) representative of a magnitude of a lamp current I_(LAMP)conducted through the lamp tube 22. A control circuit 64 is coupled tothe inverter circuit 56 for adjusting an operating frequency f_(OP) ofthe square wave voltage V_(SQ) and thus the magnitude of the lampcurrent I_(LAMP) in response to the duty-cycle DC_(PC) of thephase-control signal V_(PC) and the magnitude of the lamp currentI_(LAMP). The load regulation circuit 40 also comprises a power supply66 that receives the bus voltage V_(BUS) and generates a DC voltageV_(CC) for powering the control circuit 64.

Since the dimmer switch 30 is a two-wire dimmer switch, the compactfluorescent lamp 20 receives both power for energizing the lamp tube 22and information for determining the target intensity of the lamp tubefrom the phase-control signal V_(PC). The phase-to-DC circuit 60typically comprises a filter circuit for preventing voltage fluctuationsin the AC mains line voltage V_(AC) of the AC power source 15 or noiseon the AC mains line voltage V_(AC) from altering the magnitude of theDC voltage V_(DC) generated by the phase-to-DC converter circuit 60.Therefore, there is typically a delay time period between a change inthe duty-cycle DC_(PC) of the phase-control signal V_(PC) and aresulting change in the magnitude of the DC voltage V_(DC) generated bythe phase-to-DC converter circuit 60. If the intensity adjustmentactuator of the dimmer switch 30 is controlled such that the targetintensity is quickly reduced from the high-end intensity to the low-endintensity, the magnitude of the phase-control signal V_(PC) (and thusthe amount of power available to the load regulation circuit 40) willquickly decrease while the control circuit 64 is still controlling theintensity of the lamp tube 22 to the high-end intensity (due to thedelay time period). Accordingly, the bus capacitor C_(BUS) will quicklydischarge, such that the control circuit 64 becomes unpowered and thelamp tube 22 is extinguished, which, of course, is undesirable.

FIGS. 3A and 3B show example waveforms of the DC bus voltage V_(BUS) andthe lamp current I_(LAMP), respectively. As shown in FIG. 3A, the busvoltage V_(BUS) is characterized by some low-frequency voltage ripplehaving a frequency approximately equal to twice the frequency of the ACpower source 15 (e.g., approximately 120 Hz). The control circuit 64 istypically characterized by a corner frequency of approximately 10-20 Hz,and thus controls the inverter circuit 56 to adjust the operatingfrequency f_(OP) of the square-wave voltage V_(SQ) at a relatively slowrate in response to the lamp current control signal V_(ILAMP). Since thecorner frequency (i.e., approximately 10-20 Hz) is less than thefrequency of the voltage ripple of the bus voltage V_(BUS) (i.e.,approximately 120 Hz), the operating frequency f_(OP) of the square-wavevoltage V_(SQ) (and thus the lamp current I_(LAMP)) is maintainedrelatively constant over short time intervals (e.g., during a singlehalf-cycle T_(HC) of the AC power source 15, i.e., approximately 8.33msec). As a result, the lamp current I_(LAMP) has an envelope I_(ENV)that is characterized by the frequency of the voltage ripple of the busvoltage V_(BUS) (as shown in FIG. 3B). This fluctuation (or ripple) inthe envelope I_(ENV) of the lamp current I_(LAMP) can cause undesirableflicker in the lamp tube 22.

Accordingly, there is a need for a dimmable screw-in compact fluorescentlamp having an integral electronic ballast circuit that avoids thedisadvantages of the prior art circuits.

SUMMARY OF THE INVENTION

According to an embodiment of the present invention, a load controlcircuit controls a target intensity level of a lighting load in responseto a duty cycle of a phase-control voltage by adjusting the targetintensity level of the lighting load using a slow time constant inresponse to changes in the duty cycle of the phase-control voltage thatare below a threshold amount, and adjusts the target intensity level ofthe lighting load using a fast time constant less than the first timeconstant in response to intentional changes in the duty cycle of thephase-control voltage. The load control circuit comprises abus-voltage-generating circuit receiving the phase-control voltage andproducing a DC bus voltage, a load regulation circuit receiving said DCbus voltage and adapted to be coupled to the lighting load for adjustingthe intensity of the lighting load, and a control circuit operable todetermine a target intensity level of the lighting load in response tothe duty cycle of the phase-control voltage. The control circuitcontrols the load regulation circuit to adjust the intensity level ofthe lighting load in response to the duty cycle of the phase-controlvoltage and an actual-intensity signal related to the actual intensitylevel of the lighting load.

In addition, a phase-control-to-DC converter circuit for a load controldevice for controlling the amount of power delivered to an electricalload is also described herein. The phase-control-to-DC converter circuitreceives a phase-control voltage characterized by a duty cycle fordetermining a target amount of power to be delivered to the electricalload and generates a DC voltage representative of the duty cycle of thephase-control voltage. The phase-control-to-DC converter circuitcomprises a first filter circuit for generating a first filtered voltagein response to the duty cycle of the phase-control voltage, and a secondfilter circuit for receiving a first filtered voltage and generating asecond filtered voltage (i.e., DC voltage representative of the dutycycle of the phase-control voltage) in response to the first filteredvoltage. The first filter circuit is characterized by a first timeconstant, while the second filter circuit is characterized by a secondtime constant having a nominal value that is slower than a value of thefirst time constant of the first filter circuit. When the differencebetween the magnitudes of the first and second filtered voltages exceedsa predetermined threshold, the second time constant has a fast valuethat is faster than the nominal value, such that the magnitude of thesecond filtered voltage quickly changes to be equal to the magnitude ofthe first filtered voltage.

According to another embodiment of the present invention, a controlcircuit for a load control circuit for a lighting load receives aphase-control voltage that is characterized by a duty cycle fordetermining a target intensity level of the lighting load, and operatesto adjust said target intensity level of said lighting load using a slowtime constant in response to changes in said duty cycle of saidphase-control voltage that are below a threshold amount, and adjust saidtarget intensity level of said lighting load using a fast time constantless than the first time constant in response to intentional changes insaid duty cycle of said phase-control voltage. The control circuitcomprises a phase-control-voltage converter circuit for converting thephase-control voltage into a target signal that is used to determine thetarget intensity level of the lighting load, and an error amplifiercircuit for comparing the target signal and an actual-intensity signalrelated to the actual intensity level of the lighting load andgenerating a drive control signal for controlling said lighting load tosaid target intensity level.

According to yet another embodiment of the present invention, a lightingcontrol system receiving power from an AC power source comprises adimmable high-efficiency lighting load comprises a light source and aload regulation circuit for illuminating the light source, and atwo-wire dimmer switch adapted to be coupled in series electricalconnection between the AC power source and the dimmable compactfluorescent lamp for generating a phase-control voltage characterized bya duty cycle. The load regulation circuit receives the phase-controlvoltage and illuminates the light source in response to thephase-control voltage. The load regulation circuit comprises a controlcircuit operable to determine a target intensity level of said lamp inresponse to the duty cycle of the phase-control voltage. The controlcircuit converts the phase-control voltage into a DC target intensitysignal, and controls the load regulation circuit in response to the DCtarget intensity signal to adjust the intensity of the light source totarget intensity level. The control circuit adjusts said targetintensity level of said lamp using a slow time constant in response tochanges in said duty cycle of said phase-control voltage that are belowa threshold amount, and adjusts said target intensity level of said lampusing a fast time constant less than the first time constant in responseto intentional changes in said duty cycle of said phase-control voltage.

Further, a method of converting a duty cycle of a phase-control voltageto a DC voltage in a load control device for controlling the amount ofpower delivered to an electrical load, where the duty cycle of thephase-control voltage is used to determine a target amount of power tobe delivered to the electrical load, is also described herein. Themethod comprises the steps of: (1) generating a first filtered signal inresponse to the duty cycle of the phase-control voltage using a firsttime constant; (2) generating a second filtered signal in response tothe first filtered signal using a second nominal time constant that isslower than the first time constant when the difference between themagnitudes of the first and second filtered signals is below apredetermined threshold; and (3) generating the second filtered signalin response to the first filtered voltage using a third fast timeconstant that is faster than the nominal time constant when thedifference between the magnitudes of the first and second filteredsignals exceeds a predetermined threshold, such that the magnitude ofthe second filtered signal quickly changes to be equal to the magnitudeof the first filtered signal.

According to another embodiment of the present invention, a method ofcontrolling a compact fluorescent lamp comprises: (1) receiving aphase-control voltage characterized by a duty cycle set by a dimmerswitch that operates according to either a forward phase-control methodor reverse phase-control method; (2) determine a target intensity levelof said lamp in response to the duty cycle of the phase-control voltage;(3) providing a sinusoidal output voltage to said lamp for illuminatingsaid lamp to said target intensity level; (4) adjusting said targetintensity level of said lamp using a slow time constant in response tochanges in said duty cycle of said phase-control voltage that are belowa threshold amount; and (5) adjusting said target intensity level ofsaid lamp using a fast time constant less than the first time constantin response to intentional changes in said duty cycle of saidphase-control voltage.

Other features and advantages of the present invention will becomeapparent from the following description of the invention that refers tothe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will now be described in greater detail in the followingdetailed description with reference to the drawings in which:

FIG. 1 is a simplified block diagram of a prior art lighting controlsystem including a “two-wire” dimmer switch for controlling the amountof power delivered to a screw-in compact fluorescent lamp;

FIG. 2 is a simplified block diagram of the screw-in compact fluorescentlamp of the lighting control system of FIG. 1;

FIG. 3A shows an example waveform of a bus voltage of the screw-incompact fluorescent lamp of FIG. 2;

FIG. 3B shows an example waveform of a lamp current of the screw-incompact fluorescent lamp of FIG. 2;

FIG. 4 is a side view of a dimmable screw-in compact fluorescent lampaccording to a first embodiment of the present invention;

FIG. 5 is a simplified schematic diagram of the screw-in compactfluorescent lamp of FIG. 4;

FIG. 6 is a simplified schematic diagram of the electrical circuitry ofthe screw-in compact fluorescent lamp of FIG. 4 showing a loadregulation circuit and two power supplies in greater detail;

FIG. 7 is a simplified schematic diagram of an inverter circuit of theload regulation circuit and an over-voltage protection circuit of thescrew-in compact fluorescent lamp of FIG. 4;

FIG. 8 is a simplified schematic diagram of a control circuit of thescrew-in compact fluorescent lamp of FIG. 4;

FIG. 9 is an example plot of the magnitude of a lamp voltage withrespect to the magnitude of a lamp current of the screw-in compactfluorescent lamp of FIG. 4;

FIG. 10A shows an example waveform of a bus voltage of the screw-incompact fluorescent lamp of FIG. 4;

FIG. 10B shows an example waveform of a lamp current of the screw-incompact fluorescent lamp of FIG. 4 according to the first embodiment ofthe present invention;

FIG. 11 is a simplified schematic diagram of a control circuit of thedimmable screw-in compact fluorescent lamp according to a secondembodiment of the present invention; and

FIG. 12 is a simplified flowchart of a control procedure executedperiodically by a microprocessor of the control circuit of FIG. 11.

DETAILED DESCRIPTION OF THE INVENTION

The foregoing summary, as well as the following detailed description ofthe preferred embodiments, is better understood when read in conjunctionwith the appended drawings. For the purposes of illustrating theinvention, there is shown in the drawings an embodiment that ispresently preferred, in which like numerals represent similar partsthroughout the several views of the drawings, it being understood,however, that the invention is not limited to the specific methods andinstrumentalities disclosed.

FIG. 4 is a side view of a dimmable screw-in compact fluorescent lamp120 according to a first embodiment of the present invention. Thedimmable screw-in compact fluorescent lamp 120 comprises a light source,e.g., a fluorescent lamp tube 122, which may be formed in a spiral (asshown in FIG. 4), in two or more U-bends, or in any other suitable form.The lamp tube 122 may be filled with a fill-gas mixture having afall-gas pressure of approximately 2 Torr and a fill-gas ratio ofapproximately 85:15 argon to neon. The dimmable screw-in compactfluorescent lamp 120 further comprises an enclosure 24 for housing aload regulation circuit 130 (FIG. 5), e.g., an electronic ballastcircuit, which is electrically coupled to the lamp tube 122 forilluminating the lamp tube.

The screw-in compact fluorescent lamp 120 has a screw-in base 126adapted to be coupled to a standard Edison socket, such that the lamp isadapted to be coupled to a two-wire dimmer switch (such as the dimmerswitch 30) via the phase-control connection PC of the screw-in base 126and to the neutral side N of an AC power source via the neutralconnection NC (as in the prior art lighting control system 100 shown inFIG. 1). As defined herein, a “two-wire” dimmer switch or load controldevice does not require a direct connection to the neutral side N of theAC power source. In other words, all currents conducted by the two-wiredimmer switch must also be conducted through the load. A two-wire dimmerswitch may have only two terminals (i.e., the hot terminal H and thedimmed hot terminal DH as shown in FIG. 1). Alternatively, a two-wiredimmer switch could comprise a three-way dimmer switch that may be usedin a three-way lighting system and has at least three load terminals,but does not require a neutral connection. In addition, a two-wiredimmer switch may comprise an additional connection that provides forcommunication with a remote control device (for remotely controlling thedimmer switch), but does not require the dimmer switch to be directlyconnected to neutral.

FIG. 5 is a simplified schematic diagram of a dimmable screw-in compactfluorescent lamp 120 according to the first embodiment of the presentinvention. As previously mentioned, the screw-in base 126 provides foronly two electrical connections: the phase-control connection PC to thedimmer switch for receiving the phase-control voltage V_(PC) and theneutral connection NC to the neutral side N of the AC power source. Theload regulation circuit 130 of the screw-in compact fluorescent lamp 120is operable to adjust the intensity of the lamp tube 122 to a targetintensity L_(TRGT) (i.e., a desired intensity) in response to theduty-cycle DC_(PC) of the phase-control signal V_(PC). The targetintensity L_(TRGT) may range between a low-end intensity L_(LE) (e.g.,approximately 1%) and a high-end intensity L_(HE) (e.g., approximately100%).

The screw-in compact fluorescent lamp 120 comprises a filter network 200coupled to the phase-control connection PC and the neutral connection NCof the connector 126 for receiving the phase-control voltage V_(PC) fromthe dimmer switch 30. The filter network 200 comprises an inductor L201(e.g., having an inductance of approximately 680 μH) and two capacitorsC202, C203 (e.g., having capacitances of approximately 33 nF). Thefilter network 200 operates to prevent noise generated by the loadregulation circuit 130 from being conducted on the AC mains wiring. Thefilter network 200 couples the phase-control voltage V_(PC) to a voltagedoubler circuit 205 (i.e., a bus-voltage-generating circuit), whichgenerates a direct-current (DC) bus voltage V_(BUS) across two seriesconnected bus capacitors C_(B1), C_(B2). The first bus capacitor C_(B1)conducts the load current I_(LOAD) through a diode D206 (and the dimmerswitch 30) to charge during the positive half-cycles, while the secondbus capacitor C_(B2) conducts the load current I_(LOAD) through a diodeD208 to charge during the negative half-cycles. Accordingly, the peakmagnitude of the bus voltage V_(BUS) is approximately twice the peakvoltage of the AC mains line voltage V_(AC). A half-bus voltage V_(HB)is generated across the first bus capacitor C_(B1) and has a magnitudeequal to approximately half of the bus voltage V_(BUS).

The load regulation circuit 130 (i.e., the electronic ballast circuit)includes a half-bridge inverter circuit 210 for converting the DC busvoltage V_(BUS) to a high-frequency square-wave voltage V_(SQ) having anoperating frequency f_(OP). The load regulation circuit 130 furthercomprises an output filter circuit, e.g., a resonant tank circuit 220,for filtering the square-wave voltage V_(SQ) to produce a substantiallysinusoidal high-frequency AC voltage, which is coupled to the electrodesof the lamp tube 122. A control circuit 230 is coupled to the invertercircuit 210 for providing a drive control signal V_(DR) to the invertercircuit 210 for adjusting the operating frequency f_(OP) of the squarewave voltage V_(SQ) and thus the magnitude of a lamp current LAMPconducted through the lamp tube 122 in order to turn the lamp tube onand off and adjust the intensity of the lamp tube. Alternatively, thescrew-in compact fluorescent lamp 120 could comprise a differenthigh-efficiency lighting load, such as, a dimmable screw-in LED lightsource having an LED light engine, and the load regulation circuit 130could comprise an LED driver. An example of the LED driver 102 isdescribed in greater detail in commonly-assigned, co-pending U.S. patentapplication Ser. No. 12/813,908, filed Jun. 11, 2009, and U.S. patentapplication Ser. No. 13/416,741, filed Mar. 9, 2012, both entitled LOADCONTROL DEVICE FOR A LIGHT-EMITTING DIODE LIGHT SOURCE, the entiredisclosures of which are hereby incorporated by reference.

The screw-in compact fluorescent lamp 120 further comprises two powersupplies: an inverter power supply 240 and a control power supply 250.The inverter power supply 240 receives the half-bus voltage V_(HB)across the first bus capacitor C_(B1) and generates a direct-current(DC) inverter supply voltage V_(INV) (e.g., approximately 15 volts) forpowering the control circuitry of the inverter circuit 210. The controlpower supply 250 draws current from the resonant tank circuit 220 andgenerates a DC control supply voltage V_(CC) (e.g., approximately 12volts) for powering the control circuit 230 while the inverter circuit210 is generating the high-frequency square-wave voltage V_(SQ). Whenthe screw-in compact fluorescent lamp 120 is first powered up, theinverter power supply 240 is operable to generate the inverter supplyvoltage V_(INV) before the control power supply 250 begins to producethe control supply voltage V_(CC). After the inverter power supply 240is generating the inverter supply voltage V_(INV), the inverter circuit210 is able to begin generating the high-frequency square-wave voltageV_(SQ), such that the control power supply 250 is able to draw currentfrom the resonant tank circuit 220. Accordingly, the control powersupply 250 then produces the control supply voltage V_(CC) to power thecontrol circuit 230.

The screw-in compact fluorescent lamp 120 further comprises anover-voltage protection (OVP) circuit 260, which provides an OVP controlsignal Vow to the inverter circuit 210 for protecting the lamp tube 122and the load regulation circuit 130 during over-voltage conditions. Alamp current sense circuit 270 is coupled in series with the lamp tube122 to conduct the lamp current I_(LAMP) and to generate a lamp currentcontrol signal V_(ILAMP) representative of a magnitude of the lampcurrent I_(LAMP). The screw-in compact fluorescent lamp 120 furthercomprises a rectifier circuit 280 (e.g., a full-wave rectifier diodebridge as shown in FIG. 5), which has AC terminals coupled to receivethe phase-control voltage V_(PC) from the filter network 200 and DCterminals for providing a rectified voltage V_(RECT).

The rectified voltage V_(RECT) is received by an artificial load circuit290 for conducting an artificial load current I_(ART) through the dimmerswitch 30 in addition to the load current LOAD conducted by the buscapacitors C_(B1), C_(B2) when the bus capacitors are charging. If thedimmer switch 30 includes a triac for generating the phase-controlvoltage V_(PC), the artificial load circuit 290 is able to conductenough current to ensure that the magnitude of the total currentconducted through the triac of the dimmer switch exceeds the ratedlatching and holding currents of the triac. In addition, the artificialload circuit 290 may conduct a timing current if the dimmer switch 30comprises a timing circuit and may conduct a charging current if thedimmer switch comprises a power supply, such that these currents neednot be conducted through the load regulation circuit 130 and do notaffect the intensity of the lamp tube 122.

The artificial load circuit 290 comprises a current-passing switch,e.g., a FET Q282, coupled in series with a resistor R284 (e.g., having aresistance of approximately 12.4Ω), where the series combination of thedrain-source junction of the FET Q282 and the resistor R284 is coupledacross the DC terminals of the rectifier circuit 280. The gate of theFET Q282 is coupled to the drain of the FET Q282 through a resistor R285(e.g., having a resistance of approximately 1 Me), such that the FETQ282 is rendered conductive and conducts the artificial load currentI_(ART) when the magnitude of the rectified voltage V_(RECT) increasesfrom approximately zero volts to exceed a turn-on threshold (e.g.,approximately 4 volts). Accordingly, the FET Q282 is renderedconductive, such that the artificial load circuit 290 conducts theartificial load current I_(ART) through the dimmer switch 30 after thetriac is rendered conductive (if the dimmer switch is using the forwardphase-control dimming technique), or shortly after the beginning of eachhalf-cycle (if the dimmer switch is using the reverse phase-controldimming technique). The artificial load circuit 290 further comprises anNPN bipolar junction transistor Q286 having a base-emitter junctioncoupled across the resistor R284 and a collector coupled to the gate ofthe FET Q282. The transistor Q286 controls the FET Q282 in the linearregion to provide over-current protection of the FET Q282 when thevoltage across the resistor R284 exceeds the rated base-emitter voltageof the transistor Q286 (e.g., approximately 0.7 volt).

The artificial load circuit 290 also comprises an NPN bipolar junctiontransistor Q288 having a collector coupled to the gate of the FET Q282,and a zener diode Z292, having, for example, break-over voltage V_(BR)of approximately 13.2 volts. Three resistors R294, R295, R296 arecoupled in series between the DC terminals of the rectifier circuit 180and have, for example, resistances of approximately 22 kΩ, 470 kΩ, and150 kΩ, respectively. The zener diode Z292 is coupled between the baseof the transistor Q288 and the junction of the resistors R295, R296. Acapacitor C298 is coupled across the resistor R296 and has, for example,a capacitance of approximately 1000 pF. When the magnitude of therectified voltage V_(RECT) exceeds a turn-off threshold (e.g.,approximately 60 volts), the zener diode Z292 conducts current into thebase of the transistor Q288. Accordingly, the transistor Q288 isrendered conductive and the FET Q282 is rendered non-conductive, suchthat the artificial load circuit 290 stops conducting the artificialload current I_(ART). If the dimmer switch 30 is using the forwardphase-control dimming technique, the capacitor C298 provides some delay,such that the artificial load circuit 290 conducts the artificial loadcurrent I_(ART) for a period of time after the triac is renderedconductive.

The artificial load circuit 290 also provides a phase-control inputcontrol signal V_(PC-IN) to the control circuit 230. Specifically, theartificial load circuit 290 comprises a PNP bipolar junction transistorQ299 having a collector coupled to the control circuit 230 for providingthe phase-control input control signal V_(PC-IN). The emitter-basejunction of the transistor Q299 is coupled across the resistor R294,such that the transistor Q299 is rendered conductive shortly after thetriac is rendered conductive (with forward phase-control dimming) orshortly after the beginning of each half-cycle (with reversephase-control dimming). The control circuit 230 uses the phase-controlinput control signal V_(PC-IN) to determine the duty-cycle DC_(PC) ofthe phase-control voltage V_(PC) (i.e., the conduction time T_(CON) ofthe bidirectional semiconductor switch 32 of the dimmer switch 30). Thecontrol circuit 230 determines the present magnitude of the lamp currentI_(LAMP) from the lamp current control signal V_(ILAMP) received fromthe lamp current sense circuit 270. The control circuit 230 then adjuststhe operating frequency f_(OP) of the square wave voltage V_(SQ) tocontrol the magnitude of the lamp current I_(LAMP) to a desired levelthat is dependent upon the duty-cycle DC_(PC) of the phase-controlvoltage V_(PC) to thus control the intensity of the lamp tube 122 to thetarget intensity L_(TRGT).

FIG. 6 is a simplified schematic diagram of the electrical circuitry ofthe screw-in compact fluorescent lamp 120 showing the inverter circuit210, the resonant tank circuit 220, the inverter power supply 240, thecontrol power supply 250, and lamp current sense circuit 270 in greaterdetail. The inverter circuit 210 comprises first and secondseries-connected switching devices (e.g., FETs Q212, Q214) and aninverter control circuit 216, which may comprise, for example, aninverter control integrated circuit (IC) U300 (FIG. 5), such as partnumber FAN7532, manufactured by Fairchild Semiconductor Incorporated.The inverter control IC U300 of the inverter control circuit 216 ispowered by the inverter supply voltage V_(INV) and controls the FETsQ212, Q214 in response to the drive control signal V_(DR) received fromthe control circuit 230 as will be described in greater detail belowwith reference to FIG. 7. The inverter control circuit 216 renders theFETs Q212, Q214 conductive and non-conductive on a complementary basis(such that only one of the FETs is conducting at a time) using aconstant duty cycle (e.g., approximately 50%). When the first FET Q212is conductive, the output of the inverter circuit 210 is pulled uptowards the bus voltage V_(BUS). When the second FET Q214 is conductive,the output of the inverter circuit 210 is pulled down towards circuitcommon. The magnitude of the lamp current I_(LAMP) conducted through thelamp tube 122 is controlled by adjusting the operating frequency f_(OP)of the high-frequency square wave voltage V_(SQ) generated by theinverter circuit 210.

The resonant tank circuit 220 comprises a resonant inductor L222 adaptedto be coupled in series between the inverter circuit 210 and the lamptube 122, and a resonant capacitor C224 adapted to be coupled inparallel with the lamp tube. For example, the inductor L222 may have aninductance of approximately 1.4 mH, while the resonant capacitor C224may have a capacitance of approximately 1.2 nF, such that resonant tankcircuit is characterized by a resonant frequency of approximately 110kHz. The resonant tank circuit 220 further comprises a DC-blockingcapacitor C226 that has a capacitance of, for example, approximately 2.7nF and operates to minimize the DC component of the lamp currentI_(LAMP) conducted through the lamp tube 122.

When the lamp tube 122 is not illuminated and the control circuit 230receives a command to turn the lamp tube on (from the phase-controlvoltage V_(PC)), the control circuit first preheats filaments 228A, 228Bof the lamp tube 122 and then attempts to strike the lamp tube. The loadregulation circuit 130 may comprise, for example, two filament windings229A, 229B that are magnetically coupled to the resonant inductor L222and electrically coupled to the respective filaments 228A, 228B forgenerating filament voltages for heating the filaments of the lamp tube122 prior to striking the lamp. To preheat the filaments 228A, 228B, theinverter circuit 210 controls the operating frequency f_(OP) of thesquare wave voltage V_(SQ) to a preheat frequency f_(PH) (e.g.,approximately 150 kHz) for a preheat time period T_(PH) (e.g.,approximately 700 msec). An example of a ballast having a circuit forheating the filaments of a fluorescent lamp is described in greaterdetail in U.S. Pat. No. 7,586,268, issued Sep. 8, 2009, titled APPARATUSAND METHOD FOR CONTROLLING THE FILAMENT VOLTAGE IN AN ELECTRONIC DIMMINGBALLAST, the entire disclosure of which is hereby incorporated byreference.

The inverter power supply 240 receives the half-bus voltage V_(HB)developed across the first bus capacitor C_(B1) and generates theinverter supply voltage V_(INV) across a storage capacitor C242 (e.g.,having a capacitance of approximately 1 μF). The inverter power supply240 comprises a simple zener-regulated power supply having a zener diodeZ243, which is coupled across the storage capacitor C242 and may have,for example, a break-over voltage of approximately 13.2 volts. When thescrew-in compact fluorescent lamp 120 is first powered up, the storagecapacitor C242 is able to charge by conducting a small trickle currentfrom the AC power source 15 through a resistor R244 (e.g., having aresistance of approximately 392 kΩ) until the inverter supply voltageV_(INV) is developed across the storage capacitor C242. After themagnitude of the inverter supply voltage V_(INV) exceeds the ratedoperating voltage of the inverter control IC U300 of the invertercontrol circuit 216, the inverter control IC begins to control the FETsQ212, Q214 to be conductive and non-conductive to generate the squarewave voltage V_(SQ).

The inverter power supply 240 further comprises a snubber capacitor C245that has, for example, a capacitance of approximately 470 pF andprovides a path for charging the storage capacitor C242 after theinverter control circuit 216 begins generating the square wave voltageV_(SQ). The snubber capacitor C245 is coupled between junction of thetwo FETs Q212, Q214 and the storage capacitor C242 through a diode D246and a resistor R246 (e.g., having a resistance of approximately 5.6Ω).The storage capacitor C242 is able to charge when the first FET Q212 isconductive (i.e., when the square-wave voltage V_(SQ) is being pulledhigh towards the bus voltage V_(BUS)). When the second FET Q214 isconductive and the square-wave voltage V_(SQ) is being pulled lowtowards circuit common, the snubber capacitor C245 is able to dischargethrough the second FET Q214 and a diode D248. Accordingly, after theinverter control circuit 216 begins generating the square wave voltageV_(SQ), the inverter power supply 240 is operable to generate theinverter supply voltage V_(INV) by conducting current through thesnubber capacitor C245 rather than conducting current through theresistor R244, which would needlessly dissipate an excessive amount ofpower.

The control power supply 250 comprises a linear regulator, for example,an adjustable linear regulator U252, such as part number LM317L,manufactured by Fairchild Semiconductor Incorporated. The control powersupply 250 comprises a winding 254 magnetically coupled to the resonantinductor L222 of the resonant tank circuit 220 for generating anelectromagnetically-coupled voltage, such that the linear regulator U252is able to draw current from the resonant tank circuit through a diodeD255 when the inverter control circuit 216 is generating the square-wavevoltage V_(SQ). A capacitor C256 is coupled across the input of thelinear regulator U252 and has, for example, a capacitance ofapproximately 0.1 μF. A first resistor R258 is coupled between theoutput of the adjustable linear regulator U252 and the adjustment pin ofthe linear regulator, while a second resistor R259 is coupled betweenthe adjustment pin and circuit common. For example, the first and secondresistors R258, R259 may have resistances of approximately 475Ω and 5.23kΩ, respectively, such that the control supply voltage V_(CC) at theoutput of the linear regulator U252 has a nominal magnitude ofapproximately 15 volts.

FIG. 7 is a simplified schematic diagram of the inverter circuit 210 andthe OVP circuit 260. As previously mentioned, the inverter controlcircuit 216 includes the inverter control IC U300, which is powered bythe inverter supply voltage V_(INV). The inverter control IC U300 isdirectly coupled to the gates of the FETs Q212, Q214 for controlling theFETs to be conductive and non-conductive (via pins 16 and 13). Theinverter control circuit 216 comprises a preheat-frequency-set resistorR_(PH), which is coupled to a preheat-frequency-set input (pin 7) of theinverter control IC U300 for setting the preheat frequency f_(PH). Thepreheat-frequency-set resistor R_(PH) may have a resistance of, forexample, approximately 27 kΩ, such that the preheat frequency f_(PH) isapproximately 150 kHz. The inverter control circuit 216 also comprises apreheat-time-set capacitor C_(PH), which is coupled to apreheat-time-set input (pin 5) of the inverter control IC U300 forsetting the length of the preheat time period T_(PH). For example, thepreheat-time-set capacitor C_(PH) may have a capacitance ofapproximately 0.33 μF, such that the preheat time period T_(PH) isapproximately 700 msec.

During the preheat time period T_(PH), a voltage V_(RPH) generatedacross the preheat-frequency-set resistor R_(PH) (i.e., at pin 7) ismaintained constant, while a voltage V_(CPH) generated across thepreheat-time-set capacitor C_(PH) (i.e., at pin 5) increases inmagnitude with respect to time from approximately zero volts. When thevoltage V_(CPH) across the preheat-time-set capacitor C_(PH) exceeds apreheat voltage threshold V_(PH) at the end of the preheat time periodT_(PH), the inverter control IC U300 then controls the operatingfrequency f_(OP) to attempt to strike the lamp tube 122. The voltageV_(RPH) across the preheat-frequency-set resistor R_(PH) and the voltageV_(CPH) across the preheat-time-set capacitor C_(PH) are also providedto the control circuit 230, such that the control circuit is operable toproperly control the inverter control IC U300 during the preheat timeperiod T_(PH) as will be described in greater detail below.

The inverter control circuit 216 comprises an operating-frequency-setresistor R_(T) coupled to an frequency-set-resistor input (pin 8) of theinverter control IC U300 and an operating-frequency-set capacitor C_(T)coupled to a frequency-set-capacitor input (pin 6) for setting theoperating frequency f_(OP) of the square-wave voltage V_(SQ) when thelamp tube 122 is illuminated (i.e., after the lamp tube has beenstruck). For example, the operating-frequency-set resistor R_(T) mayhave a resistance of approximately 30 kΩ and the operating-frequency-setcapacitor C_(T) may have a capacitance of approximately 330 μF, suchthat a default operating frequency of the square-wave voltage V_(SQ) isapproximately 110 kHz.

The inverter control circuit 216 further comprises an NPN bipolarjunction transistor Q310 having a collector-emitter junction coupledbetween the frequency-set-resistor input of the inverter control IC U300and circuit common through a resistor R312 (e.g., having a resistance ofapproximately 10 kΩ). The base of the transistor Q310 is coupled toreceive the drive control signal V_(DR) from the control circuit 230.The drive control signal V_(DR) has a DC magnitude that isrepresentative of a target operating frequency f_(TRGT) to which theoperating frequency f_(OP) should be controlled to control the intensityof the lamp tube 122 to the target intensity L_(TRGT). The transistorQ310 is controlled to operate in the linear region, such that thetransistor Q310 provides a controllable impedance between thefrequency-set-resistor input of the inverter control IC U300 and circuitcommon in response to the DC magnitude of the drive control signalV_(DR). Accordingly, the control circuit 230 is operable to adjust theoperating frequency f_(OP) of the square-wave voltage V_(SQ) bycontrolling the impedance provided by the transistor Q310 and thus thecurrent conducted through the frequency-set-resistor input of theinverter control IC U300.

The inverter control circuit 216 receives the OVP control signal V_(OVP)from the OVP circuit 260. Specifically, the OVP control signal V_(OVP)is coupled to an open lamp protection (OLP) input (pin 10) of theinverter control IC U300 through a resistor R320 (e.g., having aresistance of approximately 10 kΩ), and is coupled to circuit commonthrough a capacitor C322 (e.g., having a capacitance of approximately0.1 μF). The OVP circuit 260 comprises a voltage divider havingresistors R261, R262 for scaling the magnitude of the lamp voltageV_(LAMP) down to a magnitude that is appropriate to be received by theinverter control IC U300. For example, the resistors R261, R262 may haveresistances of approximately 1 MΩ and 25.5 kΩ, respectively. Thejunction of the resistors R261, R262 is coupled to a capacitor C264(e.g., having a capacitance of approximately 0.1 μF) through a diodeD265. The junction of the capacitor C264 and the diode D265 is coupledto the inverter control circuit 216 through a zener diode Z266 forgenerating the OVP control signal V_(OVP), which is coupled to circuitcommon through a resistor R268 (e.g., having a resistance ofapproximately 100 kΩ). For example, the zener diode Z266 may have abreak-over voltage of approximately 13.2 volts.

During normal operation, the voltage at the OLP input of the invertercontrol IC U300 remains low (i.e., at approximately circuit common).However, in the event of an overvoltage condition across the lamp tube122, the zener diode Z266 begins to conduct, such that the voltage atthe OLP input of the inverter control IC U300 increases in magnitude.When the magnitude of the voltage at the OLP input exceeds an OLPthreshold voltage of the inverter control IC U300 (e.g., approximately 2volts), the inverter control IC disables the outputs (i.e., pins 13 and16) such that the FETs Q212, Q214 are rendered non-conductive and thelamp tube 122 is not illuminated until the control circuit 230 controlsthe inverter circuit 210 to attempt to restrike the lamp tube onceagain.

FIG. 8 is a simplified schematic diagram of the control circuit 230. Thecontrol circuit 230 comprises a two-speed phase-to-DC converter circuit400 for converting the phase-control input control signal V_(PC-IN) to aDC target voltage V_(TRGT) that is representative of the duty-cycleDC_(PC) of the phase-control voltage V_(PG) and thus the targetintensity L_(TRGT) of the lamp tube 122. The DC target voltage V_(TRGT)is amplified by a non-inverting amplifier circuit 420 to generate anamplified target voltage V_(A-TRGT), such that the magnitude of theamplified target voltage V_(A-TRGT) is within a correct range to bereceived by an error amplifier circuit 430. For example, the DC targetvoltage V_(TRGT) may be in the range of approximately 1-4 volts, whilethe amplified target voltage V_(A-TRGT) is in the range of approximately0.5-6.5 volts. A non-linear amplifier circuit 440 receives the lampcurrent control signal V_(ILAMP) from the lamp current sense circuit 270and generates an amplified lamp current signal V_(A-ILAMP). The erroramplifier circuit 430 receives the amplified target voltage V_(A-TRGT)and the amplified lamp current signal V_(T-ILAMP) and generates thedrive control signal V_(DR) to adjust the operating frequency f_(OP) ofthe inverter circuit 210, so as to minimize the error between theamplified target voltage V_(A-TRGT) and the amplified lamp currentsignal V_(T-ILAMP).

The phase-to-DC converter circuit 400 comprises a voltage divider havingtwo resistors R401, R402 for scaling down the phase-control inputcontrol signal V_(PC-IN). For example, the resistors R401, R402 may haveresistances of approximately 1 MΩ and 47 kΩ, respectively. Next, thephase-to-DC converter circuit 400 generates a switched voltage V_(S)that has a duty-cycle approximately equal to the duty cycle DC_(PC) ofthe phase-control input control signal V_(PC-IN). Specifically, thejunction of the resistors R401, R402 is coupled to the base of a firstNPN bipolar junction transistor Q404 that has a collector-emitterjunction coupled between the control supply voltage V_(CC) and circuitcommon through a resistor R405 (e.g., having a resistance ofapproximately 220 kΩ). The junction of the collector of the transistorQ404 and the resistor R405 is coupled to the base of a second NPNbipolar junction transistor Q406 that has a collector-emitter junctioncoupled between the control supply voltage V_(CC) and circuit commonthrough two resistors R408, R409 (e.g., having resistances ofapproximately 40 kΩ and 1 kΩ, respectively). Accordingly, the switchedvoltage V_(S) is generated at the collector of the second transistorQ406.

When the magnitude of the phase-control input control signal V_(PC-IN)is approximately zero volts (i.e., when the bidirectional semiconductorswitch 32 of the dimmer switch 30 is non-conductive), the firsttransistor Q404 is rendered non-conductive, such that the secondtransistor Q406 is rendered conductive and the switched voltage V_(S) ispulled low towards circuit common. When the magnitude of thephase-control input control signal V_(PC-IN) is greater than aphase-control threshold, e.g., approximately 15 volts (i.e., when thebidirectional semiconductor switch 32 of the dimmer switch 30 isconductive), the first transistor Q404 is rendered conductive, such thatthe second transistor Q406 is rendered non-conductive and the switchedvoltage V_(S) is pulled high towards control supply voltage V_(CC)through the resistors R408, R409.

A ramp voltage V_(R) is generated across a capacitor C410 (e.g., havinga capacitance of approximately 0.22 μF) in response to the square-wavevoltage V_(S). When the second transistor Q406 is non-conductive, thecapacitor C410 is able to charge towards the control supply voltageV_(CC) through the resistor R408, such that the magnitude of the rampvoltage V_(R) increases with respect to time while the switched voltageV_(S) is high. When the second transistor Q406 is conductive, thecapacitor C410 is able to discharge through the resistor R409, such thatthe magnitude of the ramp voltage V_(R) decreases at a second rate thatis much faster than the first rate at which the ramp voltage increasesin magnitude. Accordingly, the ramp voltage V_(R) is generated acrossthe capacitor C410 and has a duty cycle equal to approximately the dutycycle of the phase-control input control signal V_(PC-IN). Next, theramp voltage V_(R) is filtered by a first filter circuit, e.g., aresistor-capacitor (RC) circuit (including a resistor R411 and acapacitor C412), to generate a filtered voltage V_(F). For example, theresistor R411 has a resistance of approximately 220 kΩ and the capacitorC412 has a capacitance of approximately 0.22 μF, such that the first RCcircuit has a time constant τ_(RC1) of approximately 48.4 msec.

The filtered voltage V_(F) from the first RC circuit is then filtered bya second RC circuit (having a resistor R413 and a capacitor C414) togenerate the target voltage V_(TRGT). Two diodes D416, D418 are coupledin anti-parallel connection across the resistor R413. For example, theresistor R413 has a resistance of approximately 2.2 MΩ and the capacitorC414 has a capacitance of approximately 0.22 μF, such that a timeconstant τ_(RC2) of the second RC circuit has a nominal value ofapproximately 484 msec (i.e., approximately 10 times slower than thefirst RC circuit). The magnitude of the target voltage V_(TRGT) is afunction of the square of the conduction time T_(CON) of thebidirectional semiconductor switch 32 of the dimmer switch 30, i.e.,V_(TRGT) ⁼f (T_(CON) ²). Accordingly, the control circuit 230 isoperable to adjust the intensity of the lamp tube 122 in response to theduty-cycle DC_(PC) of the phase-control voltage V_(PC) according to a“square-law” dimming curve. As a result, the control circuit 230provides finer tuning of the intensity of the lamp tube 122 near thelow-end intensity L_(LE), such that larger variations in the conductiontime T_(CON) of the bidirectional semiconductor switch 32 result insmaller changes in the intensity of the lamp tube 122 near the low-endintensity L_(LE).

Voltage fluctuations in the AC mains line voltage V_(AC) of the AC powersource 15 or noise on the AC mains line voltage V_(AC) can cause theduty-cycle DC_(PC) of the phase-control signal V_(PC) and the magnitudeof the filtered voltage V_(F) to vary slightly. Therefore, the slownominal value τ_(RC2-NOM) of the time constant τ_(RC2) of the second RCcircuit provides enough filtering such that the target voltage V_(TRGT)is not responsive to changes in the filtered voltage V_(F) that are lessthan a predetermined threshold, e.g., approximately the forward voltageof the diodes D416, D418, i.e., a diode drop (e.g., approximately 0.7volts).

However, changes in the target intensity L_(TRGT) at the dimmer switch30 that result in the dimmer switch changing the duty cycle DC_(PC) ofthe phase-control signal V_(PC) cause the magnitude of the filteredvoltage V_(F) to change by greater amounts and at faster rates than thevoltage fluctuations and noise of the AC mains line voltage V_(AC).Therefore, when the magnitude of the filtered voltage V_(F) changes bymore than the forward voltage of the diodes D416, D418, one of thediodes D416, D418 will begin to conduct such that the magnitude of thetarget voltage V_(TRGT) across the capacitor C414 changes quickly inresponse to changes in the target intensity L_(TRGT). Specifically, thefirst diode D416 is operable to conduct current into the capacitor C414when the target intensity L_(TRGT) increases, such that the timeconstant τ_(RC2) has a fast value τ_(RC2-FAST) that is less than thenominal value τ_(RC2-NOM). In addition, the capacitor C414 is operableto discharge through the diode D418 with the fast value τ_(RC2-FAST)when the target intensity L_(TRGT) decreases. Accordingly, the two-speedphase-to-DC converter circuit 400 is able to filter out changes in theduty cycle DC_(PC) of the phase-control voltage V_(PC) due to voltagefluctuations and noise of the AC mains line voltage V_(AC) while stillproviding a fast response as a result of changes in the target intensityL_(TRGT).

Each of the diodes D416, D418 stops conducting when the differencebetween the magnitudes of the target voltage V_(TRGT) and the filteredvoltage V_(F) falls below approximately the forward voltage of therespective diode. After the diodes D416, D418 stop conducting, themagnitude of the target intensity L_(TRGT) will slowly change to beequal to the magnitude of the filtered voltage V_(F) (according to thenominal time constant τ_(RC2-NOM)) until the magnitudes of the voltagesare equal. This results in a slow fading of the intensity of the lamptube 122 at the end of a change in the target intensity L_(TRG), whichprovides a pleasant, soft effect on a human eye that is observing thechange in the intensity of the lamp tube.

The non-inverting amplifier circuit 420 comprises an operationalamplifier (“op amp”) U421, such as, for example, part number LM2902PWR,manufactured by National Semiconductor Corporation. The target voltageV_(TRGT) is coupled to the non-inverting input of the op amp U421, whilean offset voltage V_(OFF) is coupled to the inverting input of the opamp through a resistor R422 (e.g., having a resistance of approximately160 kΩ). The offset voltage V_(OFF) is generated by a voltage dividerthat is coupled between the control supply voltage V_(CC) and circuitcommon, and includes two resistors R424, R425. For example, theresistors R424, R425 may have resistances of approximately 33 kΩ and 5kΩ, respectively, such that the offset voltage V_(OFF) has a magnitudeof approximately 2 volts. The output of the op amp U421 is coupled theinverting input via the parallel combination of a resistor R426 (e.g.,having a resistance of approximately 150 kΩ) and a capacitor C428 (e.g.,having a capacitance of approximately 0.22 μF). The magnitude of theamplified target voltage V_(A-TRGT) ranges from approximately 0.5 to 6.5volts as the magnitude of the target voltage V_(TRGT) ranges fromapproximately 1 to 4 volts.

The non-linear amplifier circuit 440 receives the lamp current controlsignal V_(ILAMP) from the lamp current sense circuit 270, which is shownin FIG. 6. The lamp current control signal V_(ILAMP) is generated acrossthe parallel combination of a resistor R272 (e.g., having a resistanceof approximately 4Ω) and a capacitor C274 (e.g., having a capacitance ofapproximately 2.2 μF). During the negative half-cycles of the lampcurrent I_(LAMP), the lamp current is conducted through a diode D276 andthe parallel combination of the resistor R272 and the capacitor C274.During the positive half-cycles of the lamp current I_(LAMP), the lampcurrent is conducted through a diode D278 and is not conducted throughthe parallel combination of the resistor R272 and the capacitor C274.Accordingly, the lamp current control signal V_(ILAMP) has a negativemagnitude that is representative of the magnitude of the lamp currentI_(LAMP) during the negative half-cycles of the lamp current. Since theresistor R272 only conducts the lamp current I_(LAMP) every otherhalf-cycle, the resistor R272 dissipates half of the amount of powerthat would be dissipated if the resistor R272 conducted the lamp currenteach half-cycle.

Referring back to FIG. 8, the non-linear amplifier circuit 440 comprisesan op amp U441 having a non-inverting input coupled to circuit commonand an inverting input coupled to receive the lamp current controlsignal V_(ILAMP) through a resistor R442 (e.g., having a resistance ofapproximately 1 kΩ). The output of the op amp U441 is coupled to theinverting input through a resistor R444 (e.g., having a resistance ofapproximately 68.1 kΩ). The non-linear amplifier circuit 440 furthercomprises a PNP bipolar junction transistor Q445 and a resistor R446(e.g., having a resistance of approximately 27 kΩ). The seriescombination of the collector-emitter junction of the transistor Q445 andthe resistor R446 is also coupled between the inverting input and theoutput of the op amp U441. A capacitor C448 is coupled is parallel withthe resistor R446 and may have, for example, a capacitance ofapproximately 470 μF.

When the magnitude of the lamp current I_(LAMP) is less than a currentthreshold (e.g., approximately 100 mA), the magnitude of the amplifiedlamp current signal V_(A-ILAMP) is less than approximately the ratedemitter-base voltage of the transistor Q445. At this time, only theresistor R444 is coupled between the inverting input and the output ofthe op amp U441, such that the non-linear amplifier circuit 440 ischaracterized by a first gain α₁ (e.g., approximately −68). However,when the magnitude of the lamp current LAMP is greater than the currentthreshold, the transistor Q445 is rendered conductive, such that theresistor R446 is coupled in parallel with the resistor R444 between theinverting input and the output of the op amp U441. Accordingly, abovethe current threshold, the non-linear amplifier circuit 440 ischaracterized by a second gain α₂ that has a smaller magnitude than thefirst gain α₁ (e.g., approximately −25).

The error amplifier circuit 430 comprises an op amp U431 having anon-inverting input coupled to receive the amplified lamp current signalV_(A-ILAMP) and an inverting input coupled to receive the amplifiedtarget voltage V_(A-TRGT) through a resistor R432 (e.g., having aresistance of approximately 30 kΩ). The error amplifier circuit 430further comprises two capacitors C434, C435 (e.g., each having acapacitance of approximately 4.7 nF) and a resistor R436 (e.g., having aresistance of approximately 47 kΩ). The capacitor C434 is coupledbetween the inverting input and the output of the op amp U431, while theseries combination of the capacitor C435 and the resistor R436 is alsocoupled between the inverting input and the output of the op amp U431.The output of the op amp U431 is coupled to circuit common through aresister divider having two resistors R438, R439 (e.g., havingresistances of 18.7 kΩ and 6.8 kΩ, respectively), where the drivecontrol signal V_(DR) is produced at the junction of the resistors R438,R439. The error amplifier circuit 430 operates to adjust the operatingfrequency f_(OP) of the inverter circuit 210 so as to minimize the error(i.e., the difference) between the amplified lamp current signalV_(A-ILAMP) and the amplified target voltage V_(A-TRGT). For example,the error amplifier circuit 430 may be characterized by a cutofffrequency of greater than or equal to approximately 10 kHz (i.e., thecontrol circuit 230 has a bandwidth greater than or equal toapproximately 10 kHz).

As previously mentioned, the lamp tube 122 of the screw-in compactfluorescent lamp 120 of the first embodiment of the present inventionmay be filled with the fill-gas mixture having a fill-gas pressure ofapproximately 2 Torr and a fill-gas ratio of approximately 85:15 argonto neon. FIG. 9 is an example V-I curve 500 (i.e., the plot of themagnitude of the lamp voltage V_(LAMP) across the lamp tube 122 withrespect to the magnitude of the lamp current I_(LAMP) conducted throughthe lamp tube) according to the first embodiment of the presentinvention. FIG. 9 also shows an example V-I curve 550 of the prior artscrew-in compact fluorescent lamp 20 that has a fill gas of 100% argonat a pressure of approximately 4 Torr.

By combining the lamp tube 122 having the fill-gas mixture having afill-gas ratio of approximately 85:15 argon to neon at a pressure ofapproximately 2 Torr and the error amplifier 430 having a cutofffrequency of approximately 10 kHz, the V-I curve 500 of the screw-incompact fluorescent lamp 120 of the first embodiment of the presentinvention is much “flatter” than the V-I curve 550 of the prior artscrew-in compact fluorescent lamp 20 as shown in FIG. 9. In other words,the magnitude of the lamp voltage V_(LAMP) does not changes as much withrespect to the magnitude of the lamp current I_(LAMP) as in the priorart screw-in compact fluorescent lamp 20. Therefore, the size of thecomponents of the resonant tank circuit 220 may be smaller and theresonant tank circuit may operate more efficiently during normaloperation. Since the filament windings 229A, 229B are magneticallycoupled to the resonant inductor L222, the flatter V-I curve 500 of thescrew-in compact fluorescent lamp 120 of the first embodiment of thepresent invention provides for more favorable magnitudes of the filamentvoltages across the dimming range of the lamp tube 122, particularly,near the near the low-end intensity L_(LE) when the magnitude of thelamp voltage V_(LAMP) begins to decrease as the magnitude of the lampcurrent I_(LAMP) decreases.

FIGS. 10A and 10B show example waveforms of the DC bus voltage V_(BUS)and the lamp current I_(LAMP), respectively, of the screw-in compactfluorescent lamp 120 of the first embodiment of the present invention.Since the cutoff frequency of the error amplifier circuit 430 (i.e.,approximately 10 kHz) is much greater than the frequency of the voltageripple of the bus voltage V_(BUS) (i.e., approximately 120 Hz), thecontrol circuit 230 is able to adjust the operating frequency f_(OP) ofthe square-wave voltage V_(SQ) (and thus the lamp current I_(LAMP))within a single half-cycle of the AC power source 15. Therefore, theoperating frequency f_(OP) changes in magnitude in response to thevoltage ripple of the bus voltage V_(BUS) during each half-cycle and thelamp current I_(LAMP) has an envelope I_(ENV) that is relativelyconstant or flat (as shown in FIG. 10B). It has been observed that thishigh speed operation of the error amplifier circuit 430 reducesflickering in the lamp tube 122, particularly when the lamp tube is coldand/or has just been started up. In addition, since the operatingfrequency f_(OP) of the inverter circuit 210 changes during eachhalf-cycle even when the screw-in compact fluorescent lamp 120 ismaintaining the target intensity L_(TRGT) of the lamp tube 122 constant,the compact fluorescent lamp does not generate EMI noise at specificfrequencies, but spreads the frequency of the EMI noise out over arange, such that the peak magnitude of the noise is decreased.

Referring back to FIG. 8, the control circuit 230 further comprises apreheat adjustment circuit 450 that modifies the operation of thephase-to-DC converter circuit 400 while the load regulation circuit 130is preheating the filaments 228A, 228B of the lamp tube 122 (i.e.,during the preheat time period T_(PH)). The preheat adjustment circuit450 comprises an op amp U452 having a non-inverting input coupled toreceive the voltage V_(RPH) across the preheat-frequency-set resistorR_(PH) of the inverter control circuit 216, and an inverting inputcoupled to receive the voltage V_(CPH) across the preheat-time-setcapacitor C_(PH). The output of the op amp U452 is coupled to thecapacitor C410 of the phase-to-DC converter circuit 400 through aresistor R454 (e.g., having a resistance of approximately 464 kΩ). Aspreviously mentioned, while the inverter control IC U300 is preheatingthe filaments 228A, 228B of the lamp tube 122, the voltage V_(RPH)across the preheat-frequency-set resistor R_(PH) is maintained constant,while the voltage V_(CPH) across the preheat-time-set capacitor C_(PH)increases in magnitude with respect to time from approximately zerovolts. During the preheat time period T_(PH), the op amp U452 injectscurrent into the capacitor C410 of the phase-to-DC converter circuit400, such that the magnitude of the target voltage V_(TRGT) increases.Therefore, if the lamp tube 122 is being turned on to an intensity nearthe low-end intensity L_(LE), the magnitude of the lamp current I_(LAMP)will be large enough at the end of the preheat time period T_(PH) toensure that the arc current is established in the lamp tube when thelamp tube is struck. After the arc current has been properly establishedin the lamp tube 122, the op amp U452 stops injecting current into thecapacitor C410 of the phase-to-DC converter circuit 400 and theintensity of the lamp tube is controlled in response to the targetintensity L_(TRGT) determined from the duty-cycle DC_(PC) of thephase-control voltage V_(PC).

FIG. 11 is a simplified schematic diagram of a control circuit 630 ofthe dimmable screw-in compact fluorescent lamp 120 according to a secondembodiment of the present invention. The control circuit 630 comprises amicroprocessor U600 that implements the functions of the control circuit330 of the first embodiment that were executed by the two-speedphase-to-DC converter circuit 400, the non-inverting amplifier circuit420, and the error amplifier circuit 430. The control circuit 630comprises a resistive divider including two resistors R601, R602 forscaling the phase-control input control signal V_(PC-IN) to generate ascaled phase-control input control signal V_(PC-S). The microprocessorU600 receives the scaled phase-control input control signal V_(PC-S) andthe amplified target voltage V_(A-TRGT) from the non-linear amplifiercircuit 440. The microprocessor U600 generates a drive control signalV_(DR) to control the intensity level of the lamp tube 122 to a targetintensity level L_(TRGT). Alternatively, the microprocessor U600 may beimplemented as a microcontroller, a programmable logic device (PLD), anapplication specific integrated circuit (ASIC), a field-programmablegate array (FPGA), or any suitable controller or processing device.

FIG. 12 is a simplified flowchart of a control procedure 700 executedperiodically by the microprocessor U600, e.g., once every half-cycle ofthe AC power source 15. The microprocessor U600 first measures the dutycycle V_(PG) of the scaled phase-control input control signal V_(PC-S)at step 710, and then determines a requested intensity level L_(PRES)from the duty cycle V_(PG) of the scaled phase-control input controlsignal V_(PC-S) at step 712. The microprocessor U600 then determines aphase-control input change value Δ_(PC), i.e., the absolute value of thedifference between the new requested intensity level L_(PRES) and thepresent target intensity level L_(TRGT) from the last sample, at step714.

The microprocessor U600 uses a digital low-pass filter (LPF) to processthe values of the requested intensity level L_(PRES). If thephase-control input change value Δ_(PC) is less than a phase-controlinput change threshold Δ_(TH) at step 716, the microprocessor U600adjusts the pole of the digital low-pass filter such that the passbandfrequency f_(P) of the filter is a first predetermined frequency (e.g.,approximately 0.5 Hz) at step 718. If the phase-control input changevalue Δ_(PC) is not less than the phase-control input changethresholdΔTH at step 716, the microprocessor U600 adjusts the pole ofthe digital low-pass filter so that the passband frequency f_(P) of thefilter is a second predetermined frequency (e.g., approximately 5 Hz) atstep 720. Accordingly, when the duty-cycle DC_(PC) of the phase-controlsignal V_(PG) is remaining relatively constant (i.e., when thephase-control input change value Δ_(PC) is less than the phase-controlinput change threshold Δ_(TH)), the requested intensity level L_(PRES)is more heavily filtered, and the target intensity level L_(TRGT) isadjusted with a slow time constant τ_(DLPF-SLOW). When the duty cycleDC_(PC) is changing rapidly, the requested intensity level L_(PRES) isfiltered less, and the target intensity level L_(TRGT) is adjusted witha fast time constant τ_(DLPF-FAST). For example, the phase-control inputchange threshold Δ_(TH) may be equal to the target intensity levelL_(TRGT) divided by ten, such that the microprocessor U600 adjusts thetarget intensity level L_(TRGT) using the fast time constantτ_(DLPF-FAST) when there is greater than or equal to a 10% change in thetarget intensity level L_(TRGT).

After adjusting the passband frequency f_(P) of the digital low-passfilter at steps 718, 720, the microprocessor U600 processes therequested intensity level L_(PRES) through the digital low-pass filterat step 722 to determine the target intensity level L_(TRGT) to whichthe microprocessor will now control the intensity level of the lamp tube122. The microprocessor U600 then samples the amplified lamp currentsignal V_(A-ILAMP) at step 724, and determines an actual intensity levelL_(ACTUAL) from the sampled values of the amplified lamp current signalV_(A-ILAMP) at step 726. The microprocessor U600 then determines anerror e_(L), between the target intensity level L_(TRGT) and the actualintensity level L_(ACTUAL) at step 728 and controls the drive controlsignal V_(DR) in response to the error e_(L), to adjust the lightingintensity of the lamp tube 122 towards the target intensity levelL_(TRGT) at step 730 before the control procedure 700 exits.

While the present invention has been described with reference to thescrew-in compact fluorescent lamp 120, the concepts of the presentinvention could be used in lighting control systems having a ballastcircuit that is separate from the controlled fluorescent lamps, forexample, mounted to a junction box next to a lighting fixture in whichthe lamps are installed. In addition, the circuits and methods ofdetermining the target intensity level L_(TRGT) in response to the dutycycle DC_(PC) of the phase-control signal V_(PC) could be used inanother type of load control device, such as, for example, alight-emitting diode (LED) driver for driving an LED light source (i.e.,an LED light engine); a screw-in luminaire including a dimmer circuitand an incandescent or halogen lamp; a dimming circuit for controllingthe intensity of an incandescent lamp, a halogen lamp, an electroniclow-voltage lighting load, a magnetic low-voltage lighting load, oranother type of lighting load; an electronic switch, controllablecircuit breaker, or other switching device for turning electrical loadsor appliances on and off; a plug-in load control device, controllableelectrical receptacle, or controllable power strip for controlling oneor more plug-in electrical loads; and a motor control unit forcontrolling a motor load, such as a ceiling fan or an exhaust fan.

Although the present invention has been described in relation toparticular embodiments thereof, many other variations and modificationsand other uses will become apparent to those skilled in the art. It ispreferred, therefore, that the present invention be limited not by thespecific disclosure herein, but only by the appended claims.

What is claimed is:
 1. An apparatus for controlling an intensity levelof a lighting load, the apparatus configured to receive a phase-controlvoltage characterized by a duty cycle, the apparatus comprising: a loadregulation circuit configured to receive a drive control signal and tocontrol an intensity level of the lighting load in response to the drivecontrol signal; and a control circuit configured to receive a scaledversion of the phase-control voltage, and to generate the drive controlsignal to cause the load regulation circuit to adjust the intensitylevel of the lighting load towards a target level; wherein the controlcircuit is configured to: measure a duty cycle of the scaled version ofthe phase-control voltage; determine a requested level in response tothe duty cycle of the scaled version of the phase-control voltage;determine a change in the duty cycle of the scaled version of thephase-control voltage by calculating a difference between the requestedlevel and the target level; adjust the target level for the lightingload using a first time constant when the difference between therequested level and the target level is less than a threshold; andadjust the target level for the lighting load using a second timeconstant when the difference between the requested level and the targetlevel is greater than the threshold, where the second time constant isless than the first time constant.
 2. The apparatus of claim 1, whereinthe control circuit comprises a processing device configured to processthe requested level using a digital low-pass filter to determine thetarget level for the lighting load in response to the phase-controlvoltage.
 3. The apparatus of claim 2, wherein the processing device isconfigured to adjust a pole of the digital low-pass filter to set apassband frequency of the digital low-pass filter to a first frequencywhen the change in the difference between the requested level and thetarget level is less than the threshold.
 4. The apparatus of claim 3,wherein the processing device is configured to adjust the pole of thedigital low-pass filter to set the passband frequency of the digitallow-pass filter to a second frequency when the difference between therequested level and the target level is greater than the threshold. 5.The apparatus of claim 4, wherein the threshold is dependent upon thetarget level for the lighting load.
 6. The apparatus of claim 2, whereinthe processing device is configured to sample an actual-intensity signaland determine an actual level of the lighting load from theactual-intensity signal, the processing device configured to calculatean error between the actual level and the target level.
 7. The apparatusof claim 6, wherein the processing device is configured to control thedrive control signal in response to the error to adjust the intensitylevel of the lighting load towards the target level.
 8. The apparatus ofclaim 2, wherein the processing device comprises a microprocessor.
 9. Alamp configured to receive a phase-control voltage characterized by aduty cycle, the lamp comprising: a lighting load; a load regulationcircuit configured to receive a drive control signal and to control anintensity level of the lighting load in response to the drive controlsignal; and a control circuit configured to receive a scaled version ofthe phase-control voltage, and to generate the drive control signal tocause the load regulation circuit to adjust the intensity level of thelighting load towards a target level; an enclosure for housing the loadregulation circuit; and a screw-in base adapted to be coupled to astandard Edison socket; wherein the control circuit is configured to:measure a duty cycle of the scaled version of the phase-control voltage;determine a requested level in response to the duty cycle of the scaledversion of the phase-control voltage; determine a change in the dutycycle of the scaled version of the phase-control voltage by calculatinga difference between the requested level and the target level; adjustthe target level for the lighting load using a first time constant whenthe difference between the requested level and the target level is lessthan a threshold; and adjust the target level for the lighting loadusing a second time constant when the difference between the requestedlevel and the target level is greater than the threshold, where thesecond time constant is less than the first time constant.
 10. The lampof claim 9, wherein the control circuit comprises a processing deviceconfigured to process the requested level using a digital low-passfilter to determine the target level for the lighting load in responseto the phase-control voltage.
 11. The lamp of claim 10, wherein theprocessing device is configured to adjust a pole of the digital low-passfilter to set a passband frequency of the digital low-pass filter to afirst frequency when the change in the difference between the requestedlevel and the target level is less than the threshold.
 12. The lamp ofclaim 11, wherein the processing device is configured to adjust the poleof the digital low-pass filter to set the passband frequency of thedigital low-pass filter to a second frequency when the differencebetween the requested level and the target level is greater than thethreshold.
 13. The lamp of claim 12, wherein the threshold is dependentupon the target level for the lighting load.
 14. The lamp of claim 10,wherein the processing device is further configured to: sample anactual-intensity signal; determine an actual level of the lighting loadfrom the sampled actual-intensity signal; calculate an error between theactual level and the target level; and control the drive control signalin response to the error to adjust the intensity level of the lightingload towards the target level.
 15. The lamp of claim 9, wherein thelighting load comprises a LED light source and the load regulationcircuit comprises an LED driver circuit.
 16. A method of controlling anintensity level of a lighting load in response to a phase-controlvoltage, the method comprising: determining a target level for thelighting load; receiving a scaled version of the phase-control voltage;measuring a duty cycle of the scaled version of the phase-controlvoltage; determining a requested level in response to the duty cycle ofthe scaled version of the phase-control voltage; determining a change inthe duty cycle of the scaled version of the phase-control voltage bycalculating a difference between the requested level and the targetlevel; adjusting the target level for the lighting load using a firsttime constant when the difference between the requested level and thetarget level is less than a threshold; adjusting the target level forthe lighting load using a second time constant when the differencebetween the requested level and the target level is greater than thethreshold, where the second time constant is less than the first timeconstant; and controlling the intensity level of the lighting loadtowards the target level.
 17. The method of claim 16, furthercomprising: adjusting a pole of the digital low-pass filter to set apassband frequency of the digital low-pass filter in response to thecomparison between the change in the duty cycle and the threshold;wherein the passband frequency of the digital low-pass filter is set toa first frequency when the change in the duty cycle of the scaledversion of the phase-control voltage is less than the threshold, and toa second frequency when the change in the duty cycle of the scaledversion of the phase-control voltage is greater than the threshold. 18.method of claim 17, further comprising: determining the threshold basedupon the target level for the lighting load.
 19. The method of claim 16,further comprising: sample an actual-intensity signal; determining anactual level of the lighting load from the sampled actual-intensitysignal; calculating an error between the actual level and the targetlevel; and controlling the drive control signal in response to the errorto adjust the intensity level of the lighting load towards the targetlevel.
 20. The method of claim 16, further comprising: filtering outchanges in the duty cycle of the phase-control voltage that are belowthe threshold.